The invention relates to a method for damping resonant peaks at a motor star point in an electric motor which is operated using an intermediate voltage circuit converter with an input-side inductance, in particular a mains system input indicator (supply network-side input inductor,) and which, owing to characteristics of its winding sections, has a frequency response with at least one resonant frequency with respect to ground potential, and to a corresponding electric motor in which resonant peaks are damped in such a manner.
In present-day converter systems having an intermediate voltage circuit, in particular in multi-shaft converter systems of this type, system oscillations can be formed which are virtually undamped. This is especially true in converters having an intermediate voltage circuit and a regulated supply in the form of a regulated supply network-side converter, which is also referred to as an input converter.
Converters of this type are used for operating electrical machines with a variable supply frequency. Such an intermediate circuit frequency converter allows an electric motor, for example a three-phase machine such as a synchronous machine, to be operated not only in such a manner that it is linked directly to the supply network and hence has a fixed rotation speed, but also such that the fixed supply network can be replaced by an electronically produced, variable-frequency and variable-amplitude supply for powering the electrical machine.
The two supply systems, (i.e. the supply network whose amplitude and frequency are fixed, and the supply system which supplies the electrical machine with a variable amplitude and frequency), are coupled via a DC voltage storage device or a DC current storage device in the form of what is referred to as an intermediate circuit. In this case, such intermediate circuit converters essentially have three central assemblies:
a supply network-side input converter which can be designed to be unregulated (for example diode bridges) or to be regulated, in which case energy can be fed back into the supply network only by using a regulated input converter;
an energy storage device in the intermediate circuit in the form of a capacitor in the case of an intermediate voltage circuit and an inductor in the case of an intermediate current circuit; and
an output-side converter or inverter for supplying the machine, which generally uses a three-phase bridge circuit having six active current devices which can be turned off, for example IGBT transistors, to convert the DC voltage in an intermediate voltage circuit into a three-phase voltage system.
Such a converter system having an intermediate voltage circuit which, inter alia, owing to its very wide frequency and amplitude control range, is preferably used for main drives and servo drives in machine tools, robots and production machines, is shown in the illustration in FIG. 1.
The converter UR is connected to a three-phase supply network N via filter F and an energy-storage inductor whose inductance is LK. The converter UR has the described converted E, an intermediate voltage circuit with an energy-storage capacitance CZK, and an output inverter W. FIG. 1 shows a regulated converter E which is operated such that it is controlled by switching components (for example a three-phase bridge circuit composed of IGBT transistors), as a result of which the arrangement experiences excitation A1. The inverter W is likewise controlled via further switching components, for example, by means of a three-phase bridge circuit having six IGBT transistors. As a result of those switching operations the inverter W experiences excitation A2 of the system. The capacitor CZK in the intermediate voltage circuit is connected between the positive intermediate circuit rail P600 and the negative intermediate circuit rail M600. The inverter is connected on the output side via a line LT, having a protective-ground conductor PE and a shield SM, to a motor M in the form of a three-phase machine.
The fixed-frequency three-phase supply network N now supplies the intermediate circuit capacitor CZK via the input converter E and via the filter F and the energy-storage inductor LK by means of the regulated supply, with the input converter E (for example a pulse-controlled converter) operating together with the energy-storage inductor LK as a step-up converter. Once current flows into the energy-storage inductor LK, it is connected to the intermediate circuit and drives the current into the capacitor CZK. The intermediate circuit voltage may therefore be greater than the peak value of the supply network voltage.
This combination effectively represents a DC voltage source. The inverter W uses this DC voltage in the described manner to form a three-phase voltage system in which case, in contrast to the sinusoidal voltage of a three-phase generator, the output voltage does not have the profile of an ideal sinusoidal oscillation, but also has harmonics in addition to the fundamental, since it is produced electronically via a bridge circuit.
However, in addition to the described elements in such an arrangement, it is also necessary to consider parasitic capacitances which assist the formation of system oscillations in such a converter system. Thus, in addition to the filter F with the discharge capacitance CF, the input converter E, the inverter W and the motor M also have discharge capacitances CE, CW and CM to ground. Furthermore, there is also a capacitance CPE in the line LT to the protective-ground conductor PE, and a capacitance CSM in the line LT to the grounded shield SM.
It has now been found that these system oscillations are excited to a particularly pronounced extent in the converter E. Depending on the control method chosen for the supply, two or three phases of the supply network N are short-circuited, in order to pass current to the energy-storage inductor LK. If all three phases U, V, W are short-circuited, then either the positive intermediate circuit rail P600 or the negative intermediate circuit rail M600 is hard-connected to the star point of the supply network (generally close to ground potential depending on the zero phase-sequence system component). If two phases of the supply network N are short-circuited, then the relevant intermediate circuit rails P600 and M600 are hard-connected to an inductive voltage divider between the two supply network phases.
Depending on the situation relating to the supply network voltages, this voltage is in the vicinity of ground potential (approximately 50-60 V). Since the intermediate circuit capacitance CZK is generally large (continuous voltage profile), the other intermediate circuit rail is 600 V lower or higher and can thus also break down the remaining phase of the supply network. In both cases, the intermediate circuit is particularly severely deflected from its xe2x80x9cnaturalxe2x80x9d balanced steady-state position (xc2x1300 V with respect to ground), which represents a particularly severe excitation for system oscillation.
With respect to the production of undesirable system oscillations, the frequency band below 50 to 100 kHz area, which is relevant for the application, allows a resonant frequency to be calculated based on concentrated elements. In this case, the discharge capacitances CF to ground in the filter F are generally so large that they do not have a frequency-governing effect. In this case, it can be assumed that dominant excitation to oscillations takes place upstream of the described capacitances, and that the filter discharge capacitance CF can be ignored.
The resonant frequency fres(sys) of this system, which is referred to as fsys in the following text, is thus given by:
where                                           f            sys                    =                      1                          2              ⁢              π              ⁢                                                                    L                    ∑                                    ·                                      C                    ∑                                                                                      ⁢                  
                ⁢        where                            (        1        )                                          L          ∑                =                              L            K                    +                      L            F                                              (        2        )            
where LK represents the dominant component and LF the unbalanced inductive elements acting on the converter side in the filter (for example current-compensated inductors); and
Cxcexa3=CE+CW+CPE+CSM+CMxe2x80x83xe2x80x83(3) 
This relationship is shown in FIG. 2. In this case, Lxcexa3 and Cxcexa3 form a passive circuit, which is excited by excitation A and starts to oscillate at its natural resonant frequency fsys.
Accordingly, in addition to the shifts with an amplitude of 600 V, for example, that occur during operation, an additional, undesirable oscillation with an amplitude of up to several hundred volts is also modulated onto the voltages of the intermediate circuit rails P600 and M600.
In electric motors M in general, but in particular if they are designed using field coil technology (for example torque motors), a frequency response with pronounced resonant peaks with respect to ground potential can occur if such motors are excited in the common mode with respect to ground at all the motor terminals, for example due to the undesirable system oscillations described above.
These resonant points can be explained by an unbalanced equivalent circuit comprising a lattice network circuit K of parasitic elements (inductances L and discharge capacitances C) in the motor winding, as in FIG. 3 which shows the winding section of one phase U of a three-phase motor M with the three phases U, V, W whose winding sections are electrically connected to one another at the motor star point S. The input voltages of the three-phase current generated by the inverter W are applied to the outer terminals, which are opposite the star point S, of the respective winding sections.
This applies in particular to motors using field coil technology, in which individual four-pole networks in the lattice network K are possible by virtue of the construction, and essentially correspond to a single field coil. In field coil technology, the magnetic cores, which are composed of magnetic steel laminates, have teeth which act as pole cores, which are placed onto the prefabricated coils and are connected as appropriate. The individual inductances L are, as can be seen in FIG. 3, electrically connected in series, with each field coil being capacitively coupled to the pole core (magnetic steel laminate) on which the coil is fit. These respective capacitances are represented as discharge capacitances C to ground, which are formed by the magnetic core.
However, the described phenomenon can also be explained in the case of motors with a different construction (for example using what is referred to as wild winding) by a model of a lattice network K, by this model representing an equivalent circuit with identical four-pole networks in the form of LC tuned circuits, whose elements simulate the frequency response. The peak in this case occurs in the region of the star point S, which is normally not deliberately subjected to voltage loads. If the system oscillation of the converter system occurs in the vicinity of a natural motor frequency, then the insulation system to ground can be overloaded, in particular at the star point S, leading to premature failure of the motor M, since, due to resonance, considerably higher voltages can occur at the motor star point than at the motor terminals.
This is true for all voltage levels (low-voltage, medium-voltage and high-voltage systems), but particularly when the step-up converter principle is used (with the energy-storage inductor LK) on the converter side UR and a frequency response with pronounced resonant peaks with respect to ground potential occurs on the other side in the motor M as is the case in motors with a particularly low natural motor frequency because the natural damping in the motor resulting from eddy current losses and hysteresis losses etc. is particularly low.
Similar problems arise repeatedly in the field of electrical machines when transient overvoltages occur. The overvoltages are thus limited in order to avoid flashovers. For example, according to German patent document DE-A-38 26 282, a voltage-dependent metal-oxide resistor is connected in parallel with a coil in order to limit overvoltages. In German patent document DE-B-28 34 378, winding sections are short-circuited in order to damp quadrature-axis field. In a similar way, according to German patent document DE-A-24 33 618, transient overvoltages in a synchronous machine are damped by means of quadrature-axis field damper bars.
Furthermore, European patent document EP-A-0 117 764 describes how overvoltages which occur due to resonance phenomena can be suppressed by ferroelectric insulators between the coil windings. Finally, European patent document EP-B-0 681 361 addresses the problem of higher-order harmonic oscillations, which can occur in converters and rectifiers using power thyristors. The damper winding is in consequence connected to capacitors in order to form tuned circuits. The tuned circuits have a resonant frequency which is six times as high as the fundamental frequency of the synchronous machine. Higher-order harmonic oscillations on a fundamental can thus be absorbed. Nevertheless, the problem of possible resonant peaks at the star point S of a motor M still remains.
It is an object of the present invention to avoid resonant peaks excited by such system oscillations in an electric motor operated using such a converter system. This object is achieved by a method for damping resonant peaks at a motor star point in an electric motor which is operated using an intermediate voltage circuit converter with an input-side inductance, in particular a supply network-side input inductor, and which, owing to characteristics of its winding sections, has a frequency response with at least one resonant frequency with respect to ground potential, in that an impedance for damping capacitive discharge currents to ground potential, which are caused in the winding sections, is introduced into all the motor phases leading to a motor star point. For this purpose, this impedance is designed for capacitive discharge currents which are caused by system oscillations (which are excited asymmetrically with respect to ground in the motor phases) of the converter system.
It has also been found to be preferred if all the motor phases leading to a motor star point are routed through a lossy magnetic coupling core. In this case, the losses required for damping can advantageously be produced by the characteristics of the magnetic material itself. Alternatively, this can be achieved if the coupling core, for example a magnetic core, has a winding which is short-circuited via an impedance. In this case, it has been found to be advantageous, particularly with regard to possible retrofitting of motors, for all the motor phases leading to a motor star point to be routed through the lossy magnetic core at the input to the motor. If the impedance is a non-reactive resistance, then this results in a particularly simple and cost-effective implementation.
On the basis of the knowledge that each winding section of the motor forms an LC lattice network, the resistance of the non-reactive resistor is preferably determined by:       R    a    ≥            1      2        ·          1      3        ·                  L        C            
where L is the inductance and C the discharge capacitance of one lattice network element in the LC lattice network structure.
The total inductance of the coupling circuit formed with the magnetic core is preferably given by:
where,             L      H1        ≥          R              2        ⁢        π        ⁢                  xe2x80x83                ⁢                  f          0                      ,      
    ⁢  where  ,      
    ⁢            f      0        ≤                  1        2            ·              f        res              ,
with fres being a pronounced resonant frequency in the amplitude/frequency response of the motor.
In this case, the method according to the present invention can also be used for an electric motor having a number of motor star points, in particular for a linear motor or a torque motor, by carrying out the method for each motor star point, and with a suitable impedance in each case being coupled into all the motor phases leading to a motor star point. In this case as well, the impedance can be transformed in at the motor input, in which case it is then sensible to route all the motor phases to all the motor star points through a single coupling core, which must be dimensioned as appropriate. A torque motor is a machine which is designed to produce high torques, generally at low rotation speeds, for example in the form of a brushless synchronous motor with a large number of poles and permanent-magnet excitation.
If the converter is operated together with a supply network input inductor in order to provide a supply based on the step-up converter principle, then the invention results in significant advantages with regard to undamped system oscillations.
Furthermore, the object of the present invention is also achieved by an electric motor using an intermediate voltage circuit converter having an input-side inductance, in particular a supply network input inductor, having a frequency response, which is governed by characteristics of its winding sections with at least one resonant frequency with respect to ground potential, and in which all the motor phases leading to a motor star point are routed through a lossy magnetic coupling core. This can be easily achieved by using the characteristics of the magnetic material itself to produce the losses required for damping, or by the magnetic core having a winding which is short-circuited via an impedance. It preferred for the magnetic core to be arranged at the input to the motor. Since this impedance is designed for damping capacitive discharge currents with respect to ground potential which are caused by system oscillations (excited asymmetrically with respect to ground in the motor phases) of the converter system in the winding sections, system oscillations of the converter system can be suppressed particularly effectively, together with resonant peaks produced by them in the motor, on the basis of a passive circuit, particularly in conjunction with a supply network-side input inductor. It is particularly preferred for the impedance to be a non-reactive resistance. The value of this non-reactive resistance and of the total inductance is preferably governed by the same rules as for the method according to the invention, where       R    a    ≥                    1        2            ·              1        3            ·                        L          C                      ⁢          xe2x80x83        ⁢    and    ⁢          xe2x80x83        ⁢          L      H1        ≥            R              2        ⁢        π        ⁢                  xe2x80x83                ⁢                  f          0                      .  
The success of the invention can in this case be improved further by the coupling core being constructed such that it does not enter saturation at any operating point of the motor. The solution proposed by the present invention has been found to be particularly advantageous for motors with winding sections using field core technology, which each form a lattice network structure composed of inductances L and discharge capacitances C, with the impedance being used for transformer damping of these lattice network structures. This is achieved particularly well if the impedance is designed such that it damps common-mode currents, which are excited asymmetrically with respect to ground in the motor phases, of the converter system in the lattice network structure.
However, the principle of the invention can also be applied to any other desired forms of electric motors, particularly also those using what is referred to as wild winding technology, in particular low-voltage motors. This has been found to be particularly advantageous for such drives whose geometric dimensions are large and in which large slot areas result in large discharge capacitances, which lead to particularly low resonant frequencies fres. This is because the risk of resonant peaks at the motor star point is low provided such pronounced resonant points of the motor are well above any possible converter system oscillations. However, the situation changes, the closer such resonant frequencies in the frequency response of a motor with respect to ground potential are in the region of such converter system oscillations. This relates primarily to the physical size of the motor. The size of a motor governs the slot area which itself affects the capacitance CM of the motor with respect to ground potential, since this discharge capacitance increases with the size of the slot area. As the discharge capacitance CM of the motor increases, the pronounced resonant frequency fres in the amplitude/frequency response of the motor with respect to ground potential in turn falls and thus comes closer to the region of undesirable natural system frequencies fsys of the converter system. This means that, as the geometric dimensions of the motor, for example the physical length or the diameter, increase, pronounced resonant frequencies come closer to this critical region, and the problem of resonant peaks becomes more severe.
The present invention effectively counteracts this by means of the measures described above by providing a means for changing the frequency response of the motor with respect to ground potential such that there are now virtually no pronounced resonant peaks fres in the vicinity of the natural system frequencies fsys of the converter system shown in FIG. 1.